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LTC3407-3
Dual Synchronous, 1.8V/0.8A and 3.3V/0.8A 2.25MHz Step-Down DC/DC Regulator
FEATURES

DESCRIPTIO
Very Low Quiescent Current: Only 40A 1.8V at 800mA/3.3V at 800mA High Efficiency: Up to 95% 2.25MHz Constant Frequency Operation High Switch Current: 1.2A on Each Channel No Schottky Diodes Required Low RDS(ON) Internal Switches: 0.35 VIN: 3.3V to 5.5V Current Mode Operation for Excellent Line and Load Transient Response Short-Circuit Protected Low Dropout Operation: 100% Duty Cycle Ultralow Shutdown Current: IQ < 1A Power-On Reset Output Externally Synchronizable Oscillator Small Thermally Enhanced 3mm x 3mm DFN Package
The LTC(R)3407-3 is a dual, constant frequency, synchronous step down DC/DC converter. Intended for low power applications, it operates from 3.3V to 5.5V input voltage range and has a constant 2.25MHz switching frequency, allowing the use of tiny, low cost capacitors and inductors with a profile 1.0mm. Internal synchronous 0.35, 1.2A power switches provide high efficiency without the need for external Schottky diodes. A user selectable mode input is provided to allow the user to trade-off noise ripple for low power efficiency. Burst Mode(R) operation provides high efficiency at light loads, while Pulse Skip Mode provides low noise ripple at light loads. To further maximize battery life, the P-channel MOSFETs are turned on continuously in dropout (100% duty cycle), and both channels draw a total quiescent current of only 40A. In shutdown, the device draws <1A.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6580258, 6304066, 6127815, 6498466, 6611131.
APPLICATIO S

PDAs/Palmtop PCs Digital Cameras Cellular Phones Portable Media Players PC Cards Wireless and DSL Modems
TYPICAL APPLICATIO
VIN 3.3V TO 5.5V 10F RUN2 VIN
LTC3407-3 Efficiency/Power Loss
100 90 VOUT = 3.3V VOUT = 1.8V 1
RUN1 POR SW1 VOUT1
100k
EFFICIENCY (%)
80
MODE/SYNC VOUT2 3.3V 2.2H SW2 10F VOUT2 GND LTC3407-3
RESET 2.2H VOUT1 1.8V 10F
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70 60 50 40 30 20 10 1 VIN = 5V Burst Mode OPERATION NO LOAD ON OTHER CHANNEL 10 100 LOAD CURRENT (mA) VOUT = 3.3V VOUT = 1.8V 0.01
NOTE: IN DROPOUT, THE OUTPUT TRACKS THE INPUT VOLTAGE
Figure 1. 1.8V/3.3V at 800mA Step-Down Regulators
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0.1 POWER LOSS (W) 0.001 0.000 1000
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ABSOLUTE
(Note 1)
AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW VOUT1 RUN1 VIN SW1 GND 1 2 3 4 5 11 10 VOUT2 9 RUN2 8 POR 7 SW2 6 MODE/ SYNC
VIN Voltages.................................................- 0.3V to 6V VOUT1, VOUT2 Voltages .................... - 0.3V to VIN + 0.3V RUN1, RUN2 Voltages ................................ -0.3V to VIN MODE/SYNC Voltage .................................. - 0.3V to VIN SW1, SW2 Voltage ......................... - 0.3V to VIN + 0.3V POR Voltage ................................................- 0.3V to 6V Ambient Operating Temperature Range (Note 2) ....................................... - 40C to 85C Junction Temperature (Note 5) ............................. 125C Storage Temperature Range ................. - 65C to 125C
DD PACKAGE 10-LEAD (3mm x 3mm) PLASTIC DFN
TJMAX = 125C, JA = 40C/W, JC = 10C/W EXPOSED PAD (PIN 11) IS PGND, MUST BE SOLDERED TO PCB GND
ORDER PART NUMBER LTC3407EDD-3
DD PART MARKING
LCJB
Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges.
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 3.6V, unless otherwise specified. (Note 2)
SYMBOL VIN VOUT1 VOUT2 VLINE REG VLOAD REG IS PARAMETER Operating Voltage Range Output Voltage (Note 3) CONDITIONS

ELECTRICAL CHARACTERISTICS
fOSC fSYNC ILIM RDS(ON) ISW(LKG) POR
VRUN IRUN VMODE
0C TA 85C -40C TA 85C Output Voltage (Note 3) 0C TA 85C -40C TA 85C Reference Voltage Line Regulation (Note 3) VIN = 3.3V to 5.5V (Channel 1) VIN = 3.6V to 5.5V (Channel 2) Output Voltage Load Regulation (Note 3) Input DC Supply Current (Note 4) Active Mode VOUT1 = 1.5V, VOUT2 = 2.8V Sleep Mode VOUT1 = 1.9V, VOUT2 = 3.5V, MODE/SYNC = 3.6V Shutdown RUN = 0V, VIN = 5.5V, MODE/SYNC = 0V Oscillator Frequency VOUT1 = 1.8V, VOUT2 = 3.3V Synchronization Frequency Peak Switch Current Limit, Channel 1 + VIN = 3V, VOUT1 = 1.5V, VOUT2 = 2.8V, Channel 2 Duty Cycle <35% Top Switch On-Resistance (Note 6) Bottom Switch On-Resistance (Note 6) Switch Leakage Current VIN = 5V, VRUN = 0V, VOUT1 = VOUT2 = 0V Power-On Reset Threshold VOUT Ramping Down, MODE/SYNC = 0V Power-On Reset On-Resistance Power-On Reset Delay RUN Threshold RUN Leakage Current MODE Threshold Low MODE Threshold High
MIN 3.3 1.764 1.755 3.234 3.218
TYP 1.8 1.8 3.3 3.3 0.3 0.5 700 40 0.1 2.25 2.25 1.2 0.35 0.30 0.01 -8.5 100 262,144 1 0.01
MAX 5.5 1.836 1.836 3.366 3.366 0.5
UNITS V V V V V %/V %
1.8 0.95
950 60 1 2.7 1.6 0.45 0.45 1 200 1.5 1 0.5 VIN

0.3 0 VIN - 0.5
A % Cycles V A V V
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LTC3407-3
ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. No pin shall exceed 6V. Note 2: The LTC3407E-3 is guaranteed to meet specified performance from 0C to 85C. Specifications over the - 40C and 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: The LTC3407-3 is tested in a proprietary test mode that connects the output of the error amplifier to an outside servo loop. Note 4: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency. Note 5: TJ is calculated from the ambient TA and power dissipation PD according to the following formula: TJ = TA + (PD * JA). Note 6: The DFN switch on-resistance is guaranteed by correlation to wafer level measurements.
TYPICAL PERFOR A CE CHARACTERISTICS
Burst Mode Operation
SW 5V/DIV SW 5V/DIV
VOUT 20mV/DIV
IL 200mA/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 100mA 2s/DIV
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Load Step
2.5
VOUT 200mV/DIV
FREQUENCY DEVIATION (%)
IL 500mA/DIV ILOAD 500mA/DIV
FREQUENCY (MHz)
VIN = 5V 20s/DIV VOUT = 3.3V ILOAD = 80mA TO 800mA CIRCUIT OF FIGURE 1
UW
TA = 25C unless other wise specified. Load Step
Pulse Skipping Mode
VOUT 200mV/DIV IL 500mA/DIV ILOAD 500mA/DIV
VOUT 20mV/DIV
IL 200mA/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 20mA 1s/DIV
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VIN = 3.6V 20s/DIV VOUT = 1.8V ILOAD = 80mA TO 800mA CIRCUIT OF FIGURE 1
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Oscillator Frequency vs Temperature
VIN = 3.6V
Oscillator Frequency Deviation vs Supply Voltage
10 8 6 4 2 0 -2 -4 -6 -8
2.4
2.3
2.2
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2.1
2.0 -50 -25
-10
50 25 75 0 TEMPERATURE (C) 100 125
2
3
4 SUPPLY VOLTAGE (V)
5
6
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LTC3407-3 TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage vs Temperature
0.615 VIN = 3.6V 0.610 500 450 450 400 MAIN SWITCH 350 300 250 200 50 25 75 0 TEMPERATURE (C) 100 125 1 2 SYNCHRONOUS SWITCH 400 350 300 250 200 150 0.585 -50 -25 3 4 VIN (V) 5 6 7
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REFERENCE VOLTAGE (V)
RDS(ON) (m)
0.600 0.595 0.590
RDS(ON) (m)
0.605
Efficiency vs VIN
100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 2 VOUT = 1.8V Burst Mode OPERATION CH2 OFF 3 4 INPUT VOLTAGE (V)
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10mA
100mA
EFFICIENCY (%)
800mA 1mA
85 80 PULSE SKIP MODE 75 70 65 60 1 VIN = 5V, VOUT = 3.3V NO LOAD ON OTHER CHANNEL 10 100 LOAD CURRENT (mA) 1000
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EFFICIENCY (%)
Load Regulation
4 3 2
VOUT ERROR (%)
4 3 2
VOUT ERROR (%)
1 0 -1 -2 -3 -4 1 VIN = 3.6V, VOUT = 1.8V NO LOAD ON OTHER CHANNEL 10 100 LOAD CURRENT (mA) 1000
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VOUT ERROR (%)
Burst Mode OPERATION
PULSE SKIP MODE
4
UW
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TA = 25C unless other wise specified.
RDS(ON) vs Input Voltage
550 500
RDS(ON) vs Temperature
VIN = 3.6V VIN = 4.2V
100 -50 -25
MAIN SWITCH SYNCHRONOUS SWITCH 0 25 50 75 100 125 150 TEMPERATURE (C)
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Efficiency vs Load Current
100 95 Burst Mode OPERATION 90 90 100
Efficiency vs Load Current
Burst Mode OPERATION
80 PULSE SKIP MODE 70 VIN = 3.6V, VOUT = 1.8V NO LOAD ON OTHER CHANNEL 1 10 100 LOAD CURRENT (mA) 1000
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60
5
6
Load Regulation
0.80
Line Regulation
VOUT = 1.8V 0.60 IOUT = 200mA 0.40 0.20 0 -0.20 -0.40 -0.60
1 0
Burst Mode OPERATION PULSE SKIP MODE
-1 -2 -3 -4 1 VIN = 5V, VOUT = 3.3V NO LOAD ON OTHER CHANNEL 10 100 LOAD CURRENT (mA) 1000
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-0.80 -1.00 2 3 4 VIN (V)
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LTC3407-3
PI FU CTIO S
VOUT1 (Pin 1): Output Voltage Feedback Pin for Channel 1. An internal resistive divider divides the output voltage down for comparison to the internal reference voltage. RUN1 (Pin 2): Regulator 1 Enable. Forcing this pin to VIN enables regulator 1, while forcing it to GND causes regulator 1 to shut down. Do not float this pin. VIN (Pin 3): Main Power Supply. Must be closely decoupled to GND. SW1 (Pin 4): Regulator 1 Switch Node Connection to the Inductor. This pin swings from VIN to GND. GND (Pin 5): Main Ground. Connect to the (-) terminal of COUT, and (-) terminal of CIN. MODE/SYNC (Pin 6): Combination Mode Selection and Oscillator Synchronization. This pin controls the operation of the device. When tied to VIN or GND, Burst Mode operation or pulse skipping mode is selected, respectively. The oscillation frequency can be syncronized to an external oscillator applied to this pin and pulse skipping mode is automatically selected. Do not float this pin. SW2 (Pin 7): Regulator 2 Switch Node Connection to the Inductor. This pin swings from VIN to GND. POR (Pin 8): Power-On Reset . This common-drain logic output is pulled to GND when the output voltage falls below -8.5% of regulation and goes high after 262,144 clock cycles when both channels are within regulation. RUN2 (Pin 9): Regulator 2 Enable. Forcing this pin to VIN enables regulator 2, while forcing it to GND causes regulator 2 to shut down. Do not float this pin. VOUT2 (Pin 10): Output Voltage Feedback Pin for Channel 2. An internal resistive divider divides the output voltage down for comparison to the internal reference voltage. Exposed Pad (GND) (Pin 11): Power Ground. Connect to the (-) terminal of COUT, and (-) terminal of CIN. Must be connected to electrical ground on PCB.
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BLOCK DIAGRA
REGULATOR 1 MODE/SYNC 6
VOUT1
1
0.6V R1 VFB
R3 S Q RS LATCH R Q SWITCHING LOGIC AND BLANKING CIRCUIT
0.55V
0.65V
IRCMP
PGOOD1 RUN1 RUN2 2 0.6V REF 9 OSC PGOOD2 REGULATOR 2 (IDENTICAL TO REGULATOR 1) R1 = 240k, R3 = 120k FOR CHANNEL 1 R1 = 270k, R2 = 60k FOR CHANNEL 2 OSC POR COUNTER
VOUT2 10
6
-
SHUTDOWN
+
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BURST CLAMP SLOPE COMP VIN
+
EA
-
ITH 0.65V
EN SLEEP
-
ICOMP
+
5
-
+
BURST
-
UVDET
UV
+
ANTI SHOOTTHRU 4 SW1
+
OVDET OV
-
11 GND
VIN 3 VIN 8 POR
5 GND
7 SW2
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LTC3407-3
OPERATIO
The LTC3407-3 uses a constant frequency, current mode architecture. The operating frequency is set at 2.25MHz and can be synchronized to an external oscillator. Both channels share the same clock and run in-phase. To suit a variety of applications, the selectable Mode pin allows the user to choose between low noise and light load efficiency. The output voltage is set by an internal divider. An error amplfier compares the divided output voltage with a reference voltage of 0.6V and adjusts the peak inductor current accordingly. An undervoltage comparator pulls the POR output low if the output voltage is not above -8.5% of the reference voltage. The POR output will go high after 262,144 clock cycles of achieving regulation. Main Control Loop During normal operation, the top power switch (P-channel MOSFET) is turned on at the beginning of a clock cycle when the VOUT voltage is below the the regulated voltage. The current flows into the inductor and the load increases until the current limit is reached. The switch turns off and energy stored in the inductor flows through the bottom switch (N-channel MOSFET) into the load until the next clock cycle. The peak inductor current is controlled by the internally compensated ITH voltage, which is the output of the error amplifier.This amplifier compares the VFB (see Block Diagram) to the 0.6V reference. When the load current increases, the VFB voltage decreases slightly below the reference. This decrease causes the error amplifier to increase the ITH voltage until the average inductor current matches the new load current. The main control loop is shut down by pulling the RUN pin to ground. Low Current Operation Two modes are available to control the operation of the LTC3407-3 at low currents. Both modes automatically switch from continuous operation to the selected mode when the load current is low.
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To optimize efficiency, the Burst Mode operation can be selected. When the load is relatively light, the LTC3407-3 automatically switches into Burst Mode operation, in which the PMOS switch operates intermittently based on load demand with a fixed peak inductor current. By running cycles periodically, the switching losses which are dominated by the gate charge losses of the power MOSFETs are minimized. The main control loop is interrupted when the output voltage reaches the desired regulated value. A voltage comparator trips when ITH is below 0.65V, shutting off the switch and reducing the power. The output capacitor and the inductor supply the power to the load until ITH exceeds 0.65V, turning on the switch and the main control loop which starts another cycle. For lower ripple noise at low currents, the pulse skipping mode can be used. In this mode, the LTC3407-3 continues to switch at a constant frequency down to very low currents, where it will begin skipping pulses. The efficiency in pulse skip mode can be improved slightly by connecting the SW node to the MODE/SYNC input which reduces the clock frequency by approximately 30%. Dropout Operation When the input supply voltage decreases toward the output voltage, the duty cycle increases to 100% which is the dropout condition. In dropout, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drops across the internal p-channel MOSFET and the inductor. An important design consideration is that the RDS(ON) of the P-channel switch increases with decreasing input supply voltage (See Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3407-3 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information Section). Low Supply Operation To prevent unstable operation, the LTC3407-3 incorporates an Undervoltage Lockout circuit which shuts down the part when the input voltage drops below about 1.65V.
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APPLICATIO S I FOR ATIO
A general LTC3407-3 application circuit is shown in Figure 2. External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once the inductor is chosen, CIN and COUT can be selected. Inductor Selection Although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. The inductor ripple current IL decreases with higher inductance and increases with higher VIN or VOUT:
IL =
VOUT VOUT * 1- fO * L VIN
Accepting larger values of IL allows the use of low inductances, but results in higher output voltage ripple, greater core losses, and lower output current capability. A reasonable starting point for setting ripple current is IL = 0.3 * ILIM, where ILIM is the peak switch current limit. The largest ripple current IL occurs at the maximum input voltage. To guarantee that the ripple current stays below a specified maximum, the inductor value should be chosen according to the following equation:
V * 1 - OUT VIN(MAX) The inductor value will also have an effect on Burst Mode operation. The transition from low current operation begins when the peak inductor current falls below a level set by the burst clamp. Lower inductor values result in higher ripple current which causes this to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. L= VOUT fO * IL
Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characterisitics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3407-3 requires to operate.
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Table 1 shows some typical surface mount inductors that work well in LTC3407-3 applications.
Table 1. Representative Surface Mount Inductors
PART NUMBER Sumida CDRH3D16 Sumida CDRH2D11 Sumida CMD4D11 Murata LQH32CN Toko D312F Panasonic ELT5KT VALUE (H) 2.2 3.3 4.7 1.5 2.2 2.2 3.3 1.0 2.2 2.2 3.3 3.3 4.7 DCR ( MAX) 0.075 0.110 0.162 0.068 0.170 0.116 0.174 0.060 0.097 0.060 0.260 0.17 0.20 MAX DC SIZE CURRENT (A) W x L x H (mm3) 1.20 1.10 0.90 0.900 0.780 0.950 0.770 1.00 0.79 1.08 0.92 1.00 0.95 3.8 x 3.8 x 1.8 3.2 x 3.2 x 1.2 4.4 x 5.8 x 1.2 2.5 x 3.2 x 2.0 2.5 x 3.2 x 2.0 4.5 x 5.4 x 1.2
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Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT/ VIN. To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS IMAX VOUT ( VIN - VOUT ) VIN
where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM - IL/2. This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1F to 1F ceramic capacitor is also recommended on VIN for high frequency decoupling, when not using an all ceramic capacitor solution.
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LTC3407-3
APPLICATIO S I FOR ATIO
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required ESR to minimize voltage ripple and load step transients. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (VOUT) is determined by:
VOUT
1 IL ESR + 8 fO COUT
where f = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. With IL = 0.3 * ILIM the output ripple will be less than 100mV at maximum VIN and fO = 2.25MHz with: ESRCOUT < 150m Once the ESR requirements for COUT have been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement, except for an all ceramic solution. In surface mount applications, multiple capacitors may have to be paralleled to meet the capacitance, ESR or RMS current handling requirement of the application. Aluminum electrolytic, special polymer, ceramic and dry tantulum capacitors are all available in surface mount packages. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR(size) product of any aluminum electrolytic at a somewhat higher price. Special polymer capacitors, such as Sanyo POSCAP, Panasonic Special Polymer (SP), and Kemet A700, offer very low ESR, but have a lower capacitance density than other types. Tantalum capacitors have the highest capacitance density, but they have a larger ESR and it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors have a significantly larger ESR, and are often used in extremely costsensitive applications provided that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have the lowest ESR and cost, but also have the lowest capacitance density, a high voltage and temperature coefficient, and exhibit audible piezoelectric effects.
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In addition, the high Q of ceramic capacitors along with trace inductance can lead to significant ringing. In most cases, 0.1F to 1F of ceramic capacitors should also be placed close to the LTC3407-3 in parallel with the main capacitors for high frequency decoupling.
VIN 10F RUN2 VIN RUN1 POR SW1 VOUT1 GND 100k RESET L1 H VOUT1 COUT1
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MODE/SYNC L2 H VOUT2 COUT2 SW2 VOUT2 LTC3407-3
Figure 2. LTC3407-3 General Schematic
Ceramic Input and Output Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low ESR. Unfortunately, the ESR is so low that it can cause loop stability problems. Solid tantalum capacitor ESR generates a loop "zero" at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective. Also, ceramic caps are prone to temperature effects which requires the designer to check loop stability over the operating temperature range. To minimize their large temperature and voltage coefficients, only X5R or X7R ceramic capacitors should be used. A good selection of ceramic capacitors is available from Taiyo Yuden, AVX, Kemet, TDK, and Murata. Great care must be taken when using only ceramic input and output capacitors. When a ceramic capacitor is used at the input and the power is being supplied through long wires, such as from a wall adapter, a load step at the output can induce ringing at the VIN pin. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, the ringing at the input can be large enough to damage the part.
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APPLICATIO S I FOR ATIO
Since the ESR of a ceramic capacitor is so low, the input and output capacitor must instead fulfill a charge storage requirement. During a load step, the output capacitor must instantaneously supply the current to support the load until the feedback loop raises the switch current enough to support the load. The time required for the feedback loop to respond is dependent on the compensation and the output capacitor size. Typically, 3-4 cycles are required to respond to a load step, but only in the first cycle does the output drop linearly. The output droop, VDROOP, is usually about 2-3 times the linear drop of the first cycle. Thus, a good place to start is with the output capacitor size of approximately:
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COUT 2.5
IOUT fO * VDROOP
More capacitance may be required depending on the duty cycle and load step requirements. In most applications, the input capacitor is merely required to supply high frequency bypassing, since the impedance to the supply is very low. A 10F ceramic capacitor is usually enough for these conditions. Power-On Reset The POR pin is an open-drain output which pulls low when either regulator is out of regulation. When both output voltages are above -8.5% of regulation, a timer is started which releases POR after 218 clock cycles (about 117ms). This delay can be significantly longer in Burst Mode operation with low load currents, since the clock cycles only occur during a burst and there could be milliseconds of time between bursts. This can be bypassed by tying the POR output to the MODE/SYNC input, to force pulse skipping mode during a reset. In addition, if the output voltage faults during Burst Mode sleep, POR could have a slight delay for an undervoltage output condition and may not respond to an overvoltage output. This can be avoided by using pulse skipping mode instead. When either channel is shut down, the POR output is pulled low, since one or both of the channels are not in regulation. Mode Selection & Frequency Synchronization The MODE/SYNC pin is a multipurpose pin which provides mode selection and frequency synchronization.
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Connecting this pin to VIN enables Burst Mode operation, which provides the best low current efficiency at the cost of a higher output voltage ripple. Connecting this pin to ground selects pulse skipping mode, which provides the lowest output ripple, at the cost of low current efficiency. The LTC3407-3 can also be synchronized to an external 2.25MHz clock signal (such as the SW pin on another LTC3407-3)by the MODE/SYNC pin. During synchronization, the mode is set to pulse skipping and the top switch turn-on is synchronized to the rising edge of the external clock. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD * ESR, where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT, generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second-order overshoot/DC ratio cannot be used to determine phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. In some applications, a more severe transient can be caused by switching in loads with large (>1F) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem, if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot SwapTM controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting.
Hot Swap is registered trademark of Linear Technology Corporation.
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LTC3407-3
APPLICATIO S I FOR ATIO
Efficiency Considerations
The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, 4 main sources usually account for most of the losses in LTC3407-3 circuits: 1)VIN quiescent current, 2) switching losses, 3) I2R losses, 4) other losses. 1) The VIN current is the DC supply current given in the Electrical Characteristics which excludes MOSFET driver and control currents. VIN current results in a small (<0.1%) loss that increases with VIN, even at no load. 2) The switching current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the gate charges of the internal top and bottom MOSFET switches. The gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 3) I2R losses are calculated from the DC resistances of the internal switches, RSW, and external inductor, RL. In continuous mode, the average output current flows through inductor L, but is "chopped" between the internal top and bottom switches. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 - DC)
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The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT2(RSW + RL) 4) Other `hidden' losses such as copper trace and internal battery resistances can account for additional efficiency degradations in portable systems. It is very important to include these "system" level losses in the design of a system. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. Other losses including diode conduction losses during dead-time and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In a majority of applications, the LTC3407-3 does not dissipate much heat due to its high efficiency. However, in applications where the LTC3407-3 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150C, both power switches will turn off and the SW node will become high impedance. To prevent the LTC3407-3 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TRISE = PD * JA where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TRISE + TAMBIENT As an example, consider the case when the LTC3407-3 is in dropout on both channels at an input voltage of 2.7V
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LTC3407-3
APPLICATIO S I FOR ATIO
with a load current of 800mA and an ambient temperature of 70C. From the Typical Performance Characteristics graph of Switch Resistance, the RDS(ON) resistance of the main switch is 0.425. Therefore, power dissipated by each channel is: PD = I2 * RDS(ON) = 272mW The DFN package junction-to-ambient thermal resistance, JA, is 40C/W. Therefore, the junction temperature of the regulator operating in a 70C ambient temperature is approximately: TJ = 2 * 0.272 * 40 + 70 = 91.8C which is below the absolute maximum junction temperature of 125C. Design Example As a design example, consider using the LTC3407-3 in an portable application with a Li-Ion battery. The battery provides a VIN = 2.8V to 4.2V. The load requires a maximum of 800mA in active mode and 2mA in standby mode. The output voltage is VOUT = 1.8V. Since the load still needs power in standby, Burst Mode operation is selected for good low load efficiency. First, calculate the inductor value for about 30% ripple current at maximum VIN:
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1.8 V 1.8 V L= * 1- = 1.5H 2.25MHz * 300mA 4.2V
VIN CIN RUN2 L2 VOUT2 SW2 VFB2 COUT2
BOLD LINES INDICATE HIGH CURRENT PATH
Figure 3. LTC3407-3 Layout Diagram (See Board Layout Checklist)
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Choosing a vendor's closest inductor value of 2.2H, results in a maximum ripple current of:
IL = 1.8 V 1.8 V * 1- = 208mA 2.25MHz * 2.2H 4.2V
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For cost reasons, a ceramic capacitor will be used. COUT selection is then based on load step droop instead of ESR requirements. For a 5% output droop:
COUT 1.8 800mA = 7.1F 2.25MHz *(5% * 1.8 V)
A good standard value is 10F. Since the output impedance of a Li-Ion battery is very low, CIN is typically 10F. The PGOOD pin is a common drain output and requires a pull-up resistor. A 100k resistor is used for adequate speed. Figure 1 shows the complete schematic for this design example. Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3407-3. These items are also illustrated graphically in the layout diagram of Figure 3. Check the following in your layout: 1. Does the capacitor CIN connect to the power VIN (Pin 3) and GND (exposed pad) as close as possible? This capacitor provides the AC current to the internal power MOSFETs and their drivers.
VIN
RUN1 POR L1 SW1 VFB1 VOUT1
MODE/SYNC LTC3407-3
GND
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COUT1
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LTC3407-3
APPLICATIO S I FOR ATIO
2. Are the COUT and L1 closely connected? The (-) plate of COUT returns current to GND and the (-) plate of CIN. 3. The output feedback line should be routed away from noisy components and traces, such as the SW line (Pins 4 and 7), and its trace should be minimized. 4. Keep sensitive components away from the SW pins. The input capacitor CIN should be routed away from the SW traces and the inductors.
TYPICAL APPLICATIO S
Low Ripple Buck Regulators Using Ceramic Capacitors
VIN 5V
C1* 10F RUN2 L2 4.7H SW2 VOUT2 GND VIN RUN1 POR MODE/SYNC
VOUT2 3.3V 800mA C3 10F
C1, C2, C3: TAIYO YUDEN JMK316BJ106ML L1, L2: SUMIDA CDRH2D18/HP-4R7NC *IF C1 IS GREATER THAN 3" FROM POWER SOURCE, ADDITIONAL CAPACITANCE MAY BE REQUIRED.
Efficiency vs Load Current
100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 10 VIN = 5V PULSE SKIP MODE NO LOAD ON OTHER CHANNEL 100 LOAD CURRENT (mA) 1000
34073 TA02b
VOUT = 3.3V
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5. A ground plane is preferred, but if not available, keep the signal and power grounds segregated with small signal components returning to the GND pin at one point and should not share the high current path of CIN or COUT. 6. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. These copper areas should be connected to VIN or GND.
R5 100k L1 4.7H SW1 VOUT1 POWER-ON RESET VOUT1 1.8V 800mA C2 10F
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LTC3407-3
VOUT = 1.8V
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LTC3407-3
TYPICAL APPLICATIO S
1mm Height Core Supply
VIN 5V
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VOUT2 3.3V 800mA C3 10F
EFFICIENCY (%)
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C1* 10F RUN2 L2 2.2H SW2 VOUT2 GND VIN RUN1 POR MODE/SYNC LTC3407-3 SW1 VOUT1
R5 100k L1 2.2H POWER-ON RESET VOUT1 1.8V 800mA C2 10F
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C1, C2, C3: TAIYO YUDEN JMK212BJ106MD-B L1, L2: COILCRAFT LDO3310-222MX *IF C1 IS GREATER THAN 3" FROM POWER SOURCE, ADDITIONAL CAPACITANCE MAY BE REQUIRED.
Efficiency vs Load Current
100 95 90 85 80 75 70 65 60 VIN = 5V 55 Burst Mode OPERATION NO LOAD ON OTHER CHANNEL 50 10 100 1 LOAD CURRENT (mA)
VOUT = 3.3V
VOUT = 1.8V
1000
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LTC3407-3
PACKAGE DESCRIPTIO
3.50 0.05 1.65 0.05 2.15 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.38 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS R = 0.115 TYP 6 0.38 0.10 10
PIN 1 TOP MARK (SEE NOTE 6) 5 0.200 REF 0.75 0.05 2.38 0.10 (2 SIDES) 1
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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DD Package 10-Lead Plastic DFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1699)
0.675 0.05 3.00 0.10 (4 SIDES) 1.65 0.10 (2 SIDES)
(DD10) DFN 1103
0.25 0.05 0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
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TYPICAL APPLICATIO U
VIN 2.8V TO 4.2V C1 10F RUN2 VIN RUN1 POR L1 2.2H SW1 R5 100k POWER-ON RESET VOUT1 1.8V 800mA MODE/SYNC VOUT2 3.3V 200mA D1 L2 10H SW2 M1 VOUT2 GND VOUT1 LTC3407-3
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2mm Height Lithium-Ion Single Inductor Buck-Boost Regulator and a Buck Regulator
+
C6 47F C3 10F
C2 10F
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C1, C2, C3: TAIYO YUDEN JMK316BJ106ML C6: SANYO 6TPB47M D1: PHILIPS PMEG2010
L1: MURATA LQH32CN2R2M33 L2: TOKO A914BYW-100M (D52LC SERIES) M1: SILICONIX Si2302
Efficiency vs Load Current
90 80 EFFICIENCY (%) EFFICIENCY (%) 70 60 50 40 VOUT = 3.3V Burst Mode OPERATION NO LOD ON OTHER CHANNEL 30 1 10 100 LOAD CURRENT (mA) 4.2V 100 95
Efficiency vs Load Current
2.8V 90 85 80 75 70 65 60 VOUT = 1.8V Burst Mode OPERATION NO LOAD ON OTHER CHANNEL 1 10 100 LOAD CURRENT (mA) 1000
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4.2V
3.6V
3.6V 2.8V
1000
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RELATED PARTS
PART NUMBER LT1940 LTC3405/LTC3405A LTC3406/LTC3406B LT3407/LT3407-2 LTC3411 LTC3412 LTC3414 LTC3417/LTC3417A LTC3440 LTC3548/LTC3548-1 LTC3548-2 DESCRIPTION Dual Output 1.4A(IOUT), Constant 1.1MHz, High Efficiency Step-Down DC/DC Converter 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converters 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converters 600mA/800mA, 1.5MHz Dual Synchronous Step-Down DC/DC Converter 1.25A (IOUT), 4MHz, Synchronous Step Down DC/DC Converter 2.5A (IOUT), 4MHz, Synchronous Step Down DC/DC Converter 4A (IOUT), 4MHz, Synchronous Step Down DC/DC Converter 1.5A (IOUT), 4MHz, Synchronous Step Down DC/DC Converter 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 2.25MHz/400mA/800mA (IOUT) Dual Synchrnonous Step-Down DC/DC Converter COMMENTS VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = <1A, TSSOP-16E Package 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 20A, ISD <1A, ThinSOT Package 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20A, ISD <1A, ThinSOT Package 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD <1A, MSE, DFN Package 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60A, ISD <1A, MSOP-10 Package 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60A, ISD <1A, TSSOP-16E Package 95% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64A, ISD <1A, TSSOP-28E Package VIN: 2.2V to 5.5V, VOUT(MIN) = 0.8V, IQ = 125A, ISD <1A, DFN, TSSOP Packages 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 2.5V, IQ = 25A, ISD <1A, MSOP-10 Package 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40mA, ISD <1A, MSOP-10, DFN-10 Packages
34073fa LT 1006 REV A * PRINTED IN USA
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Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
FAX: (408) 434-0507 www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2006

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